1. Technical Field
The present disclosure relates to an oscillator, in particular to an oscillator operating at millimeter wave frequencies.
2. Description of the Related Art
The use of VCOs (Voltage-Controlled Oscillators) is known in the state of the art, especially in wireless telecommunications systems; said oscillators are mainly used within phase locked loops or PLLs to implement the fundamental synthesis function of the reference carrier. Designing oscillators is particularly difficult due to the ever increasing number of applications which exploit a carrier with frequencies of the order of 100 GHz, close to the transition frequencies of the most advanced technological processes on silicon.
A VCO based on an inductor-capacitor (LC) resonator is mainly used in integrated circuits for wireless radiofrequency applications, as such an oscillator is capable of providing high spectral purity while being capable of limiting consumptions. The Colpitts circuit based on an LC series resonator is a circuit topology mainly used in high frequency applications (generally higher than a few tens of GHzs); the resonator of said circuit has a lower parasitic load.
However, the Colpitts oscillator has some drawbacks. Indeed, the use of modern bipolar technologies with the miniaturization of the transistors reduces the breakdown voltages, with a consequential lowering of the power supply voltages; this results in a limitation of the amplitude and dynamics of the signals which may be generated.
The resonator quality factor in the range of millimeter waves is mainly determined by the performance of the components which provide the variable capacitance. These elements are often implemented by means of varactors, consisting of diodes or metal-oxide semiconductor transistors (MOSs). Such varactors degrade the phase noise of the VCO both due to the worsening of the same quality factor thereof, and due to the conversion of the AM-PM noise due to the non-linear voltage/capacitance feature. Using varactor elements of minimum size ensures the best performances of the device in terms of quality factor and linearity. The parasitic capacitances are needed to be reduced for maximizing the obtainable calibration range of the variable capacitance or tuning range. The tuning range specifications are highly stringent if related to the dimension of the usable varactors, as not only the variation of the reference frequency is to be ensured within the operating band, but there is a need to compensate the frequency drifts due to thermal, environmental and/or process variations.
Some voltage-controlled oscillators have a circuit configuration of differential type which has different advantages especially at high frequency, i.e., the high common mode rejection of the disturbances and the increased robustness with respect to the parasitic inductive ground and power paths on the chip or also by means of the connections made with wires. FIG. 1 shows a typical Colpitts oscillator for applications with millimeter waves. The oscillator in FIG. 1 comprises two npn bipolar transistors Q1 and Q2 having the emitter terminals connected in parallel to a series of two variable capacitors C2 and a series of two inductors Le; the common terminal of the two inductors is connected to a reference current Iee while a voltage Vtune is applied to the common terminal of the two variable capacitors C2. The series of two inductors Lb, the common terminal of which is connected to a bias voltage Vb, is arranged between the base terminals of the two transistors Q1 and Q2. There is a capacitor C1 which includes the intrinsic base-emitter capacitance Cbe and a possible external capacitance, between the emitter and base terminals of each of the transistors Q1 and Q2. The collector terminals of transistors Q1 and Q2 are connected to the power supply voltage Vcc by means of two inductors Lc; the output voltage Vo is between the collector terminals.
There is a capacitive series impedance with negative real part from the base terminals of each of the transistors Q1 and Q2. The presence of the inductive element Lb determines a resonance at the frequency
            F      0        =          1              2        ⁢        π        ⁢                              Lb            ⁢                                                  ⁢                                          C                ⁢                                                                  ⁢                1                ⁢                                                                  ⁢                C                ⁢                                                                  ⁢                2                                                              C                  ⁢                                                                          ⁢                  1                                +                                  C                  ⁢                                                                          ⁢                  2                                                                          ,and the oscillation is triggered if the negative impedance is able to compensate for the circuit leakages. The capacitors C2 are adjusted by means of the control voltage Vtune and determine the tuning range of the oscillator. The inductance Le offers high impedance between the emitter terminals at the oscillation frequency and further provides a decoupling of the variable capacitors C2 with respect to the parasitic capacitances of the current source Iee, thus preserving the tuning range. The impedance due to the element Lc on the collector provides a load by means of which the signal is provided towards the transmission/reception buffers and/or towards the dividers. An inductor Le2 may be inserted between the emitter terminal of transistors Q1 and Q2 and the inductors Le, which improves the tuning range features. Indeed, the series network consisting of the components Le2 and C2 is sized so as to offer capacitive impedance with improved performance of the quality factor and tuning range with respect to variable capacitors C2 only. In this case the quick variation of the equivalent capacitive impedance of the series consisting of components Le2 and C2 is exploited to increase the tuning range, as well as the increased quality factor due to the resonance approaching.
The greatest drawback of the oscillator in FIG. 1 is the presence of the Miller effect on the capacitance C1, i.e., the intrinsic capacitance between collector and base, of the transistors Q1 and Q2 due to the finite voltage gain between base terminals and collector. An equivalent capacitance CM of comparable or even higher value to/than the capacitors C1 and C2 is on the base terminal of each of the transistors Q1 and Q2. The capacitance CM is a fixed capacitive component parallel to the inductor Lb which introduces a severe limitation on the tuning range in applications having millimeter waves. The capacitance CM reduces the equivalent negative resistance of the active circuit, thus worsening the start-up condition. The load impedance Lc causes a signal to develop on the collector terminals at the oscillation frequency; the oscillation which develops on the base terminal is repeated on the emitter terminal and on the collector terminal with an opposite sign. The active-area operation for each transistor Q1, Q2 is ensured if vb,max−vc,min<Vcc−Vb+Vbe−Vce,sat, where vb,max and vc,min are the maximum signal voltage on the base and the minimum signal voltage on the collector, respectively, while the voltage Vbe is the bias voltage between the base terminals and emitter, and the voltage Vce,sat is the collector-emitter saturation voltage. Reduced output dynamics may limit the phase noise performances obtainable from the oscillator. The presence of the capacitance CM reduces the isolation between the collector terminal output and the resonator. A lesser isolation causes an increased influence of the load circuit on the impedance of the resonator.
Collecting the output signal from the collector terminals is not a substantial advantage at millimeter wave frequencies, as the current gain between collector and base under conditions of oscillation is usually less than one.
A cascode stage may be introduced in the circuit in FIG. 1, i.e., a stage of two npn bipolar transistors Q3 and Q4 having the emitter terminals connected with the collector terminals of the transistors Q1 and Q2, the collector terminals connected with the terminals of the inductors Lc and the base terminals connected to a bias voltage Vb1, as seen in FIG. 2. Introducing a cascode buffer in the circuit in FIG. 1 allows the capacitance CM not to be subjected to amplification due to the low input impedance of the common base stage, but there is a significant dynamics limitation introduced by the cascode configuration, as better described in the article by H. Li and H.-M. Rein, “Millimeter-Wave VCOs With Wide Tuning Range and Low Phase Noise, Fully Integrated in a SiGe Bipolar Production Technology,” IEEE J. Solid-State Circuits, vol. 38, pp. 184-191, February 2003. Such a solution is not advantageous if used on modern technological bipolar processes and/or CMOSs having a power supply voltage significantly less than 3V.
FIG. 3 illustrates a topology of a Colpitts VCO in which the capacitances Cn were inserted between the base terminal of transistor Q1 and the collector terminal of transistor Q2, and between the base terminal of transistor Q2 and the collector terminal of transistor Q1; thereby there is a compensation for the base-collector capacitance, as described in the article by S. T. Nicolson, et. al., “Design and Scaling of W-Band SiGe BiCMOS VCOs,” IEEE J. Solid-State Circuits, vol. 42, pp. 1821-1833, September 2007. However, the amplitude limits remain because of the bipolar saturation depending on the configuration with load on the collector.